Document Text (Pages 51-60) Back to Document

Integrated Circuit Interface for SAW Biosensors Applications

Page 51

Chapter 4: SAW Sensor Interface Circuit Design 38

In order to calculate the IDT capacitance, equation 3.3 is used. Substituting the
dielectric constant of lithium tantalate εp= 50 and the parameters from the table

=
×
× ×
= 50 × 8.85 × 10��� × 1440 × 10�� × 2.5 = 1.58 푝퐹
eq.4.2

The electromechanical coupling coefficient K
2 of the lithium tantalate SAWs is
between 0.01 and 0.05 (Shibata et al., 1995). So taking K
2=0.03 is a reasonable
approximation.

Substituting from table 4.1 into equation 3.6 to calculate the radiation
conductance

= 8

= 8 × 0.3 × 228.79 × 10

× 1.58 × 10

���

× 2.5

0.544 푚퐴/

eq.4.3

Equation 3.5 is used to calculate the effective conductance. There is no
information given regarding the array reflectivity Г which is kept as small as
possible. Accordingly, its effect will be very small.

The motional resistance will be calculated as shown in equation 3.4

= =

.���×����

1.8푘Ω eq.4.4

The effect of array reflectivity will reduce the motional resistance by a small
factor so it will be set to

= 1.75푘Ω

Re-writing equation 3.2 as follows
= �� �

=

���

eq.4.5

The quality factor Q of the SAW resonator is around 100, substituting in
equation 4.5 yields

=

× ×���

=

The motional inductance is given by:

���×.��×����××.��×���.��×��

4푓퐹 eq.4.6

=

��××

=

×.��×��.��×��××��

���

121.1 휇퐻 eq.4.7

The calculated SAW resonator circuit model is shown in figure 4-4

Page 52

Chapter 4: SAW Sensor Interface Circuit Design 39

Figure 4- 4 model Calculated SAW resonator circuit

The frequency and phase response of the SAW resonator model was simulated
using Cadence Virtuoso Spectre simulator. The result is shown in figure 4-5.

Figure 4- 5 SAW model simulated gain and phase response

It can be seen that the dB loss of the SAW is about 12 dB and the two ports
encounter 1800 phase shift.

Using equation 4.1 gives:

���� ���� =
��� 180° = 360° eq. 4.8

The amplifier phase shift caused by the components parasitic capacitances and
should be 1800.

Page 53

Chapter 4: SAW Sensor Interface Circuit Design 40

The required amplifier gain is at least 20dB to guarantee the start of oscillation.
So a minimum gain ratio of at least 10 must be introduced at the resonator frequency.

As discussed by Allen et al. (2002), current source inverting amplifiers have
superior gain to their active load counterparts. Push-pull CMOS inverters have
higher large-signal gains than their inverting amplifier counterparts. The main
disadvantage of a push-pull inverter is its DC stability. Since the oscillator is based
on small-signal amplification (noise or switching signal), current source amplifier
topology is used in the oscillator design.

The single stage inverting amplifier schematic is shown in figure 4-6.

Vdd

MP

Vb

Vin
MN

Figure 4- 6 Current source inverter amplifier

Source: Allen, P.E., Holberg, D.R., 2002. CMOS Analog Integrated Circuits, Oxford University
Press, Oxford, pp. 172.

An additional transistor Mb is used for biasing the NMOS (figure 4-7) and
keeps the gate and drain with the same DC voltage to maintain MN in saturation
using a very low aspect ratio (Nordin et al., 2006). The chosen (W/L)p =
(0.8µm/60µm).

A by-pass capacitor Cbp is inserted to AC couple the signal to the input gate of
the NMOS. For the operation frequency around 229MHz, the AC impedance
introduced by the by-pass capacitor is calculated as

=

��×(������)×��(��)

eq. 4.9

Xc is the impedance introduced by the by-pass capacitance.

From equation 4.9, it is clear that a capacitance in the range of 3pF will
provide negligible AC impedance at the working frequency.

Page 54

Chapter 4: SAW Sensor Interface Circuit Design 41

Figure 4- 7 Complete current inverter amplifier schematics

So the amplifier complete schematics is shown in figure 4-7

The amplifier parameters are calculated as below (Allen et al., 2002):

All n and p subscripts denote NMOS and PMOS transistors respectively.

Maximum Voltage Gain for NMOS:

��� =

gm is the trans-conductance

��

(���� ����)

= ����

� �

� �

� 훼 1/
eq. 4.10

gds is the drain- source conductance calculated as: gds= λ ID

Kp is the gain factor

λ is the velocity saturation

ID is the drain current

W and L are the transistor width and the length, respectively

The amplifier transfer function:

() =

�����

(�� ������ )
���� ����

eq.4.11

Cout is the output capacitance and ω is the angular frequency.

Page 55

Chapter 4: SAW Sensor Interface Circuit Design 42

The 3dB angular frequency:
��� = (���� ����)

���

eq.4.12

So the 3dB frequency is:
f��� = (��� � ����)
(� � ����) = (� �)

(� � ����)

eq.4.13

It can be seen from equation 4.10 that gain is inversely proportional to the DC
current and from equation 4.13 that the bandwidth is proportional to the DC current.

The current is calculated as follows:

=

� 퐾���� − 푉�� ����

(1 + ���) eq.4.14

Vb1 is the biasing voltage and Vt is the transistor threshold voltage

To ensure the PMOS works in saturation mode the following condition must
be met:

VDS - (Vdd – Vb1 – Vtp) ≥ 0.3 V

Referring to appendix A.2.1 for MOS parameters, and substituting VDSp = Vdd -Vout

Vout ≤ Vb1 + 1.4

Choosing a low biasing voltage for the PMOS increases its source-to-gate
voltage. Consequently the drain current increases, so the static power consumption
increases. Exploring the available analogue cells in the XFAB XC06 process, it can
be seen that choosing a band gap cell (abgpc01) with an output of 2.6444V
(appendix A.3.1) is a convenient choice to have a reasonable bandwidth and avoid
high power consumption. The band-gap cell output voltage typical temperature
coefficient is 50ppm which increases the biasing voltage by 4mV when temperature
increases by 30°C.

From equation 4.11, it can be seen that the bandwidth is inversely proportional
to Cout.

Since the amplifier will be used in the oscillator closed loop, the gate-source
capacitance Cgs will be the output capacitance of the preceding stage Cout.

For a transistor working in saturation Cgs = 2/3 WL Cox

Cox is the oxide capacitance per unit area

Page 56

Chapter 4: SAW Sensor Interface Circuit Design 43

Choosing the amplifier NMOS transistor dimensions should be taken carefully.
The length is taken equal to the minimum process dimension (0.6 µm). Increasing
width will increase gain but also decrease bandwidth so choosing Wn = 120 µm will
give a reasonable gain.

Using appendix A.2.1, and substituting in equation 4.10 yields

��� =

��.����� (��)

Choosing ID with a value of 2mA will degrade the maximum gain and increase
the power consumption. Decreasing the drain current below 1mA limits the amplifier
bandwidth to the order of tens of MHz, so ID will be set arbitrarily to 1.2mA.

���

��.����� (��)

��.�����
√�.22 eq. 4.15

This gain will be degraded according to the bandwidth limitation, moreover,
there will be also loading effect due to the non-ideal AC coupling at the amplifier
input, so one stage of amplification will not be sufficient for starting oscillation.

Substituting ID = 1.2mA into equation 4.14 yields:

(
)=

.

×.���×(���.������.)

38 eq. 4.16

Taking Lp equals to the minimum process dimension 0.6 µm, so

Wp= 38 × 0.6 =22 µm

So the PMOS width and length will be set Wp and Lp will be 21µm and 0.6µm,
respectively.

So calculating the drain current using equation 4.14 taking the chosen
dimensions into considerations results in:

=

� 퐾���� − 푉�� ����

1 + ���× �� × 0.040 ×

� �.

(5 2.6444 1.1)1.1 푚퐴 eq. 4.17

Calculating 3-dB bandwidth using equation 4.13
f��� = (��� � ����)

(� � ����) = (� �)
(� � (����)

(.��������.������)∗�.

×.��×
×.��×����×���×.)

95MHz

eq. 4.18

The designed amplifier transfer function was simulated using Cadence
Virtuoso Spectre simulator. The result is shown in figure 4-8.

Page 57

Chapter 4: SAW Sensor Interface Circuit Design 44

Figure 4- 8 Amplifier transfer function

The open loop phase shift is shown below

= 180° 180°< 푆퐴푊 − 휃
− 휃
eq. 4.19

θ1 and θ2 are the phase shifts of the first and second amplifier stages,

respectively.

The SAW resonator phase shift changes around the series resonance frequency,
so along with θ1 and θ2 due to parasitic capacitances and resistances, a total of (-360
º) can be obtained.

= arctan ( ) eq. 4.20

���

A buffer amplifier is required to extract the output signal for further
processing.

The buffer amplifier is composed of a CMOS amplifier with a biased PMOS
and the input feeds the NMOS (Nordin et al., 2006). The NMOS (W/L) ratio is set to
a high aspect ratio (120µm/0.6µm) to permit a good amplification. The PMOS aspect

Page 58

Chapter 4: SAW Sensor Interface Circuit Design 45

ratio is set to (60µm/3µm) to allow the oscillator to have a DC signal output which
will be used to bias the mixer later. The buffer schematic diagram is shown in figure
4-9.

V dd

V bias V in

Figure 4- 9 Buffer schematic diagram

The complete oscillator block diagram is shown in figure 4-10.

Output

Amplifier 1 Amplifier 2 Buffer
SAW resonator

Figure 4- 10 Oscillator block diagram

The oscillator frequency is not exactly the resonant frequency of the SAW
resonator; the parallel SAW IDT and parasitic capacitances of the amplifier and the
low pass filter add a slight parallel capacitance to the motional capacitance. So the
oscillator frequency will be slightly lower than that of the resonator. The parasitic
capacitance is fixed and very small compared to the motional capacitance.

The oscillator frequency fosc is calculated as follows:

��� =

��

=

�� (������)

eq. 4.21

Lm and Cm are the motional inductance and capacitance, respectively.

Ct is the total effective capacitance.

Page 59

Chapter 4: SAW Sensor Interface Circuit Design 46

Cpar is the total parallel capacitance.

Re-writing equation 4.21 gives

��� = 1

2π L(C+ C���) = 1

2π LC
( 1
1 + (C���/C))

��� =

��

1 + (C���/C)��/eq. 4.22

For Cpar << Cm equation 4.22 can be reduced to:

��� =

��
1 + (C���/C)��/=

��

(1

���

× )

��� = (1 0.5 ���) eq.4.23

fm is the SAW resonator motional (series) resonance frequency =

��

4.3 Mixer

The mixer is essential to the design because the high frequency operation makes the
absolute frequency measurement inaccurate. In our application the difference
frequency is only interesting without paying attention to the signal amplitude so an
unbalanced mixer will be used. The square-law mixer schematic is shown in figure
4-11 (Liu et al., 2006).

Vdd

Sensing oscillator Output

Reference Oscillator

Figure 4- 11 Square law mixer

Source: Liu, L., Wang, Z. 2006. Analysis and Design of a low-voltage RF CMOS Mixer, IEEE
Transactions on Circuits and Systems, vol. 53, no. 3, pp. 212-216

Page 60

Chapter 4: SAW Sensor Interface Circuit Design 47

The oscillator output has a DC voltage level, so instead of filtering the
oscillator DC signal; the DC level will be used to bias both transistors.

The voltage output can be written as in Liu et al. (2006) and replacing the local
oscillator signal vLO by vsens and RF signal vRF by vref :

��� =
��

µn
(�� � ��������� )
µp
����� ����������� ��� ��

µn
(�� ���� )���

(

��� ����������� ��� )

µn
(�� ���� )��� ����

(��� ����������� ��� )

eq.4.24

vsens and vref are the sensing and reference oscillator AC voltages, respectively

Vdc is the DC level of the oscillator signal.

µ is the transistor mobility.

It can be seen from the last term that the vsens and vref are multiplied so mixing
is achieved (Liu et al., 2006).

Placing a low-pass filter composed of two RC sections will filter all unwanted
signals such as harmonics and harmonic signals of the multiplier. The mixer
difference frequency is expected to be less than 2.5MHz so the filter cut-off
frequency fcut-off is set to 2.8MHz. Choosing the resistance R equals to 12kΩ, the

filter capacitance =

�� � ��������

= 4.7푝퐹.

So the final mixer schematic is shown in figure 4-12.

Figure 4- 12 Mixer schematics